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TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 1 copyright ? 2008 trinamic motion control gmbh & co. kg trinamic ? motion control gmbh & co. kg sternstra?e 67 d C 20357 hamburg germany www.trinamic.com 1 features the TMC603 is a three phase motor driver for highly compact and energy efficient drive solutions. i t contains all power and analog circuitry required for a high performance bldc motor system. the TMC603 is designed to provide the frontend for a microcontroller doing motor commutation and control algorithms. it directly drives 6 external n - channel mosfet s for motor currents up to 30a and up to 50v and integrates shunt less current measurement , by using the mosfets channel resistance for sensing . integrated h all fx (pat . fil.) allows for sensorless commutation. protection and diagnostic features as well as a step down switching regulator further reduce system cost and increase reliability. highlights up to 3 0a m otor current 9v to 50v operating voltage 3.3v or 5v interface 8mm x 8mm qfn package integrated shunt less c urrent m easurement using p ower mos t rans istor rdson h all fx ? s ensorless back emf commutation emulates h all sensors integrated b reak - bef ore - m ake logic : no s pecial microcontroller pwm hardware required emv optimiz ed current controlled gate drivers C up to 150ma possible overcurrent / s hort to gnd a nd undervoltage protection and diagnostics integrated internal q gd protection: s upport s latest generation of p ower mosfets integrated supply concept: s tep down switching regulator up to 500ma / 300khz common rail charge pump allows for 100% pwm duty cycle applications motor driver for industrial applications integrated miniaturized drives robotics high - reliability drives (dual position sensor possible) pump and blower applications with sensorless commutation motor type 3 phase bldc , s tepper , dc m otor sin e or block commutation rotor position feedback: sensorless, encoder or hall sensor, or any mix TMC603 C datasheet three phase motor driver with bldc back emf commutation h all fx ? and current sensing
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 2 copyright ? 2008 trinamic motion control gmbh & co. kg life support policy trinamic motion control gmbh & co. kg does not authorize or warrant any of its products for use in life support systems, without the spe cific written consent of trinamic motion control gmbh & co. kg. life support systems are equipment intended to support or sustain life, and whose failure to perform, when properly used in accordance with instructions provided, can be reasonably expected to result in personal injury or death. ? trinamic motion control gmbh & co. kg 2008 information given in this data sheet is believed to be accurate and reliable. however no responsibility is assumed for the consequences of its use nor for any infringement of patents or other rights of third parties which may result fr o m its use. specifications subject to change without notice
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 3 copyright ? 2008 trinamic motion control gmbh & co. kg 2 table of contents 1 features ................................ ................................ ................................ ................................ .......... 1 2 table of contents ................................ ................................ ................................ ........................ 3 3 system architecture using the TMC603 ................................ ................................ ................... 5 4 pinout ................................ ................................ ................................ ................................ ............. 6 4.1 p ackage codes ................................ ................................ ................................ ............................. 6 4.2 p ackage dimensions qfn52 ................................ ................................ ................................ ........... 7 5 TMC603 functional bl ocks ................................ ................................ ................................ .......... 8 5.1 b lock diagram and pin description ................................ ................................ ................................ 8 5.2 mosfet d river s tage ................................ ................................ ................................ ................ 10 5.2.1 principle of operation ................................ ................................ ................................ ....... 10 5.2.2 break - before - make logic ................................ ................................ ................................ ... 11 5.2.3 pwm control via microcontroller ................................ ................................ ...................... 12 5.2.4 slope control ................................ ................................ ................................ .................... 13 5.2.5 reverse capacity (qgd) protection ................................ ................................ .................... 14 5.2.6 considerations for qgd protection ................................ ................................ ................... 15 5.2.7 effects of the mosfet bulk diode ................................ ................................ ..................... 16 5.2.8 adding schottky diodes across the mosfet bulk diodes ................................ ................. 16 5.2.9 short to gnd detection ................................ ................................ ................................ .... 17 5.2.10 error logic ................................ ................................ ................................ ......................... 17 5.3 c urrent measurement a mplifiers ................................ ................................ ................................ .. 19 5.3.1 current measurement timing ................................ ................................ ............................ 19 5.3.2 auto zero cycle ................................ ................................ ................................ ................. 20 5.3.3 measurement dep ending on chopper cycle ................................ ................................ ...... 20 5.3.4 compensating for offset voltages ................................ ................................ .................... 20 5.3.5 getting a precise current value ................................ ................................ ........................ 20 5.4 hall fx sensorless commutati on ................................ ................................ ................................ .. 21 5.4.1 adjusting the hallfx spike suppression time ................................ ................................ ... 21 5.4.2 a djusting the hallfx filter frequency ................................ ................................ ................ 22 5.4.3 block commutation chopper scheme for hallfx ................................ ............................... 22 5.4.4 start - up sequence for the motor wit h forced commutation ................................ ............ 23 5.5 p ower supply ................................ ................................ ................................ ............................. 25 5.5.1 switching regulator and charge pump ................................ ................................ ............. 25 5.5.2 charge pump ................................ ................................ ................................ .................... 27 5.5.3 supply voltage filtering ................................ ................................ ................................ .... 27 5.5.4 reverse polarity protection ................................ ................................ ............................... 27 5.5.5 standby with zero power consumption ................................ ................................ ........... 27 5.5.6 low voltage operation down to 9v ................................ ................................ ................. 28 5.6 t est out put ................................ ................................ ................................ ................................ 29 6 absolute maximum rat ings ................................ ................................ ................................ ..... 30 7 electrical character istics ................................ ................................ ................................ ...... 30 7.1 o perational r ange ................................ ................................ ................................ ..................... 30 7.2 dc c haracteristics and t iming c haracteristics ................................ ................................ ........... 31 8 designing the applic ation ................................ ................................ ................................ ...... 38 8.1 c hoosing the best fit ting power mosfet ................................ ................................ .................... 38 8.1.1 calculating the mosfet power dissipation ................................ ................................ ...... 38 8.2 mo sfet examples ................................ ................................ ................................ ....................... 40 8.3 d riving a dc motor with the TMC603 ................................ ................................ ......................... 40 9 table of figures ................................ ................................ ................................ ......................... 41
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 4 copyright ? 2008 trinamic motion control gmbh & co. kg 10 revision history ................................ ................................ ................................ .................... 41 10.1 d ocumentation r evision ................................ ................................ ................................ ......... 41
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 5 copyright ? 2008 trinamic motion control gmbh & co. kg 3 system a rchitecture using the TMC603 figure 1 : application block diagram the TMC603 is a bldc driver ic using external power mos transistor s . its unique feature set allows equipping inexpensive and small drive systems with a maximum of intelligence , protecti on and diagnostic features. control algor ithms previously only found in much more complex servo drives can now be realized with a minimum of external components. depending on the desired commutation scheme and the bus interface requirements , the TMC603 forms a co mplete motor driver system in comb ination with an external 8 bit processor or with a more powerful 32 b it processor . a simple system can work with three standard pwm outputs even for sine commutation! the complete analog amplification and filtering frontend as well as the power driver cont roller are realized in the TMC603. its integrated support for sine commutation as well as for back emf sensing save s cost and allow s for maximum drive efficiency. the external microcontroller realizes commutation and control algorithms. based on the posi tion information from an encoder or hall sensors, the microcontroller can do block commutation or sine commutation with or without space vector modulation and realizes control algorithms like a pid regulator for velocity or position or field oriented contr ol based on the current signals from the TMC603. for sensorless commutation, the microcontroller needs to do a forward controlled motor start without feedback. this can be realized either using block commutation or sine commutation . a sine commutated start - up minimize s motor vibrations during start up. as soon as the minimum velocity for h all fx is reached, it can switch to block commutation and drive the motor based on the h all fx signals. the TMC603 also supports control of three phase stepper motors as w ell as two phase stepper motors using two devices. h s - d r i v e l s - d r i v e + v m h s l s 1 o f 3 s h o w n 5 v l i n e a r r e g u l a t o r 1 2 v s t e p d o w n r e g u l a t o r b r e a k b e f o r e m a k e l o g i c s l o p e c o n t r o l s l o p e h s s l o p e l s g a t e o f f d e t e c t i o n e r r o r l o g i c s h o r t t o g n d 1 , 2 , 3 b r i d g e c u r r e n t m e a s u r e m e n t s h o r t t o g n d d e t e c t i o n h a l l f x t m f o r s e n s o r l e s s c o m m u t a t i o n d r i v e r s e c t i o n n f e t p o w e r m o s h a l f b r i d g e s b l d c m o t o r n s t m c 6 0 3 m i c r o c o n t r o l l e r p o w e r b u s / i o p o s i t i o n s e n s o r
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 6 copyright ? 2008 trinamic motion control gmbh & co. kg 4 pinout figure 2 : pinning / qfn52 package (top view) 4.1 package codes type package temperature range code/marking TMC603 qfn52 (rohs) - 4 0c .. . +1 25 c TMC603 - la t m c 6 0 3 - l a q f n 5 2 8 m m x 8 m m 0 . 5 p i t c h / e r r _ o u t t e s t h s 1 g n d p l s 1 b m 2 l s 2 b m 1 v c p h s 2 v l s h 2 g n d p c u r 1 b h 1 b l 1 b h 3 b h 2 c u r 2 b l 2 s a m p l e 2 1 v l s h 3 f i l t 1 h 1 r s 2 g g n d v m e n a b l e i n v _ b l c l r _ e r r r s l p g n d s w o u t v c p s c c l k c o s c f i l t 3 f i l t 2 b l 3 s a m p l e 3 c u r 3 5 v o u t s p _ s u p v c c s e n s e _ h i b b m _ e n h s 3 g n d p b m 3 l s 3 s a m p l e 1 2 3 4 5 6 7 8 9 1 0 1 1 1 2 1 3 1 4 1 5 1 6 1 7 1 8 1 9 2 0 2 1 2 2 2 3 2 4 2 5 2 6 3 9 3 8 3 7 3 6 3 5 3 4 3 3 3 2 3 1 3 0 2 9 2 8 2 7 5 2 5 1 5 0 4 9 4 8 4 7 4 6 4 5 4 4 4 3 4 2 4 1 4 0
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 7 copyright ? 2008 trinamic motion control gmbh & co. kg 4.2 package dimensions qfn52 ref min nom max a 0.80 0. 85 0 . 9 0 a1 0.00 0.0 35 0.05 a2 - 0.65 0.67 a3 0.20 3 b 0.2 0.25 0.3 d 8 .0 e 8 .0 e 0.5 j 6 . 1 6 . 2 6 . 3 k 6.1 6.2 6.3 l 0. 3 5 0. 4 0. 4 5 all dimensions are in mm. attention: dra wing not to scale.
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 8 copyright ? 2008 trinamic motion control gmbh & co. kg 5 TMC603 functional blocks 5.1 block diagram and pin description figure 3 : application diagram the application diagram shows the basic building blocks of the ic and the connections to the power bridge transistors, as well as the power supply. the connection of t he digital and analog i/o lines to the microcontroller is highly specific to the microcontroller model used. + v m swout hs - drive vcp ls - drive + v m vls bh 1 hs 1 ls 1 bm 1 1 of 3 shown 5 v linear regulator 5 vout 100 nf vm vcc gnd 220 r 100 vcp rslp bav 99 ( 70 v ) bas 40 - 04 w ( 40 v ) ss 16 tp 0610 k or bss 84 ( opt . bc 857 ) l sw 12 v step down regulator vls break before make logic gndp slope control slope hs slope ls opt . for high qgd fets : mss 1 p 3 / zhcs 1000 gate off detection bl 1 d d inv _ bl bbm _ en d / err _ out set reset error logic short to gnd 1 , 2 , 3 clr _ err d undervoltage vls , vcp r ds current sense ls short to gnd detection rs 2 g short to gnd 1 bm 1 bm 1 track & hold stage sample 1 cur 1 bridge current measurement motor coil output switched capacitor filter sensorless commutation driver section bm 1 bm 2 bm 3 scclk d cosc d sense _ hi hall sensor emulation d h 1 d h 2 d h 3 filt 1 filt 2 filt 3 a a a test logic test d d d a d enable 1 of 3 power mos half bridges vcc signed current , centered at 1 / 3 vcc automatic sample point delay sp _ sup 100 n ( 2 x ) 220 n c osc : 470 p - > 100 khz r slp : 100 k - > 100 ma c sup : 1 n - > 90 s r s 2 g : 470 k - > 2000 ns 100 n ( 2 x ) provide sufficient filtering capacity near bridge transistors ( electrolyt capacitors and ceramic capacitors ) l sw : 220 h for 100 khz amplification 4 . 5 x or 18 x v m + 10 v charge pump 5 v supply 12 v supply ( 150 ma with sel . transistor ) 4 7 tantal 25 v 220 n 16 v zener 12 v bzt 52 b 12 - v / bzv 55 c 12 rs 2 g , rslp and bmx : use short trace and avoid stray capacitance to switching signals . place resistors near pin . tmc 603 gnd die pad
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 9 copyright ? 2008 trinamic motion control gmbh & co. kg pin number type function vls 1, 44 low side driver supply voltage for driving low side gates gndp 2, 40, 52 power gnd for mosfet drivers, connect directly to gnd vm 3 motor and mosfet bridge suppl y voltage gnd 4, 36 digital and analog low power gnd rs2g 5 ai 5v short to gnd control resistor. controls delay time for short to gnd test hx 6, 7, 8 do 5v h allfx outputs for back emf based hall sensor emulation filtx 9, 10, 11 ao 5v output of interna l switched capacitor filter cosc 12 a 5v oscillator capacitor for step down regulator scclk 13 di 5v switched ca pacitor filter clock input for h allfx filters. bhx 14, 18, 22 di 5v high side driver control signal: a positive level switches on the high si de blx 15, 19, 23 di 5v low side driver control signal: polarity can be reversed via inv_bl samplex 16, 20, 24 di 5v optional external control for current measurement sample/hold stage. set to positive level, if unused curx 17, 21, 25 ao 5v output of cu rrent measurement amplifier 5vout 26 output of internal 5v linear regulator. provided for vcc supply sp_sup 27 a 5v an external capacitor on this pin controls the commutation spike suppression tim e for h allfx. vcc 28 +5v supply input for digital i/os and analog circuitry sense_hi 29 di 5v switches current amplifiers to high sensitivity bbm_en 30 di 5v enables internal break - before - make circuitry inv_bl 31 di 5v allows inversion of blx input active level (low: blx is active high) enable 32 di 5v ena bles the power drivers (low: all mosfets become actively switched off ) /err_out 33 do 5v error output (open drain) . signals undervoltage or overcurrent. tie to enable for direct self protection of the driver clr_err 34 di 5v reset of error flip - flop (act ive high). clears error condition rslp 35 ai 5v slope control resistor. sets output current for mosfet drivers swout 37 o switch regulator transistor output test 38 o 12v test multiplexer output (leave open) vcp 39 charge pump supply voltage. provides high side driver supply lsx 41, 45, 49 o 12v low side mosfet driver output bmx 42, 46, 50 i (vm) sensing input for bridge outputs. used for mosfet control and current measurement. hsx 43, 47, 51 o (vcp) high side mosfet driver output exposed die pad - gnd connect the exposed die pad to a gnd plane. it is used for cooling of the ic and may either be left open or be connected to gnd.
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 10 copyright ? 2008 trinamic motion control gmbh & co. kg 5.2 mosfet driver stage the TMC603 provides three half bridge drivers, each capable of driving two mosfet transistors, o ne for the high - side and one for the low - side. in order to provide a low rdson, the mosfet gate driving voltage is about 10v to 12v. the TMC603 bridge drivers provide a number of unique features for simple operation , explained in the following chapters : an integrated automatic break - before - make logic safely switches off one transistor before its counterpart can be switched on. slope controlled operation allows adaptation of the driver strength to the desired slope and to the chosen transistors. the dri vers protect the bridge actively against cross conduction (q gd protection) the bridge is protected against a short to gnd figure 4 : t hree phase bldc driver 5.2.1 principle of operation the low side gate driver voltage is supplied by the vls pins. the low side driver supplies 0v to the mosfet gate to close the mosfet, and vls to open it. the tmc 603 uses a patented driver principle for driving of the high side: the high - side mosfet gate voltage is referenced to its source at the center of the half bridge. due to this, the TMC603 references the gate drive to the bridge center (bm) and has to be ab le to drive it to a voltage lying above the positive bridge power supply voltage vm. this is realized by a charge pump voltage generated from the switching regulator via a villard circuit. when closing the high - side mosfet, the high - side driver drives it d own to the actual bm potential, since an external induction current from the motor coil could force the output to stay at high potential. this is accomplished by a feedback loop and transistor tg1 (see figure). in order to avoid floating of the output bm, a low current is still fed into the hs output via transistor tg1a. th e input bm helps the high side driver to track the bridge voltage. since input pins of the TMC603 must not go below - 0.7v, the input bm needs to be protected by a n external resistor. th e resistor limits the current into bm to a level, the esd protection input diodes can accept. figure 5 : pr inciple of high - side driver (pat.fil.) tmc 603 3 phase bldc motor hs 3 bm 3 ls 3 220 r hs - drv ls - drv hs - drv ls - drv hs - drv ls - drv + v m z 12 v hs 2 bm 2 ls 2 220 r + v m z 12 v hs 1 bm 1 ls 1 220 r gndp + v m z 12 v vcp vls high side driver one coil of motor one nmos halfbridge hs on hs off tg 1 tg 1 a t 1 hs bm ls 220 r + v m z 12 v i on vcp i off i holdoff
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 11 copyright ? 2008 trinamic motion control gmbh & co. kg a zener diode at the gate (range 12v to 15v) protects the high - side mosfet in case of a short to gnd event: should the bridge be shorted, the gate driver output is forced to stay at a maximum of the zener voltage abo ve the source of the transistor. further it prevents the gate voltage from dropping below source level. the maximum permissible mosfet driver current depe nds on the motor supply voltage: parameter symbol max unit mosfet driver current with v vm < 30v i h sx , i lsx 150 ma mosfet driver current with 30v < v vm < 50v 150 - 2.5*( v vm - 30v) ma mosfet driver current with v vm = 50v i hsx , i lsx 100 ma pin comments lsx low side mosfet driver output. the driver current is set by resistor r slp hsx high side mosfet d river output. the driver current is set by resistor r slp bmx bridge center used for current sensing and for control of the high side driver. for unused bridges, connect bmx pin to gnd and leave the driver outputs unconnected. place the external protecti on resistor near the ic pin. rslp the resistor connected to this pin controls the mosfet gate driver current. a 4 0a current out of this pin (resistor value of 100 k to gnd) results in the nominal maximum current at full supply range. keep interconnection between ic and resistor short, to avoid stray capacitance to adjacent signal traces of modulating the set current. resistor range: 60 k to 500 k vls low side driver supply voltage for driving low side gates vcp charge pump supply voltage. provides hig h side driver supply gndp power gnd for mosfet drivers, connect directly to gnd bhx high side driver control signal: a positive level switches on the high side . for unused bridges, tie to gnd. blx low side driver control signal: polarity can be reverse d via inv_bl inv_bl allows inversion of blx input active level (low: blx is active high) . when high, e ach blx and bhx can be connected in parallel in order to use only 3 pwm outputs for bridge control. be sure to switch on internal b reak - b efore - m ake logic (bbm_en = vcc) to avoid bridge short circuits in this case . 5.2.2 break - before - make logic each half - bridge has to be protected against cross conduction during switching events. when switching off the low - side mosfet, its gate first needs to be discharged, be fore the high side mosfet is allowed to be switched on. the same goes when switching off the high - side mosfet and switching on the low - side mosfet. the time for charging and discharging of the mosfet gates depends on the mosfet gate charge and the driver c urrent set b y r slp . when the bbm logic is enabled, the TMC603 measures the gate voltage and automatically delays switching on of the opposite bridge transistor, until its counterpart is discharged. the bbm logic also prevents unintentional bridge short cir cuits, in case both, lsx and hsx, become switched on. the first active signal has priority. alternatively, t he required time can be calculated and pre - compensated in the pwm block of the microcontroller driving the TMC603 (external bbm control) .
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 12 copyright ? 2008 trinamic motion control gmbh & co. kg figure 6 : bridge driver timing pin comments bbm_en enables internal break - before - make circuitry (high = enable) 5.2.3 pwm control via microcontroller there is a number of different microcontrollers available, which provide specific bldc commutation units. however, the TMC603 is designed in a way in order to allow bldc control via standard microcontrollers, which have onl y a limited number of (free) pwm units. the following figure shows several possibilities to control the bldc motor with different types of microcontrollers, and shall help to optimally adapt the TMC603 control interface to the features of your microcontrol ler. the hall signals and further signals, like curx interconnection to an adc input , are not shown. blx bhx c o n t r o l s i g n a l s m o s f e t d r i v e r s lsx bmx hsx 0 v v vls 0 v 0 v 0 v 0 v v vm v vcp internal bbm control external bbm control v vm load pulling bmx down load pulling bmx up to + vm t lson t lsoff t bbmlh t bbmhl miller plateau hsx - bmx 0 v v vcp - v vm miller plateau t hson
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 13 copyright ? 2008 trinamic motion control gmbh & co. kg figure 7 : examples for microcontroller pwm control 5.2.4 s lope control the TMC603 driver stage provides a constant current output stage slope control. this allows to adapt driver strength to the drive requirements of the power mosfet and to adjust the output slope by providing for a controlled gate charge and discharge. a slower slope causes less electromagnetic emission, but at the same time power dissipation of the power transistors rises. the duration of the complete switching event depends on the to tal gate charge. the voltage transition of the output takes place during the so called miller plateau (see figure 6 ). the miller plateau results from the gate to drain capacity of the mosfet charging / discharging during the switc hing. from the datasheet of the transistor (see example in figure 8 ) it can be seen, that the miller plateau typically covers only a part (e.g. one quarter) of the complete charging event. the gate voltage level, where the miller plateau starts, depends on the gate threshold voltage of the transistor and on the actual load current. figure 8 : mosfet gate charge as available in device data sheet vs. switching event t m c 6 0 3 p w m 1 o u t 1 m i c r o c o n t r o l l e r w i t h b l d c p w m u n i t b h 1 b l 1 p w m 1 o u t 2 t m c 6 0 3 p w m 1 o u t m i c r o c o n t r o l l e r w i t h 3 p w m o u t p u t s b h 1 b l 1 i n v _ b l b b m _ e n + v c c b l o c k ( h a l l o r h a l l f x ) o r s i n e c o m m u t a t e d b l d c m o t o r s i n e c o m m u t a t e d b l d c m o t o r t m c 6 0 3 p w m 1 o u t m i c r o c o n t r o l l e r w i t h 3 p w m o u t p u t s b h 1 b l 1 b l o c k ( h a l l ) c o m m u t a t e d b l d c m o t o r d i g o u t t m c 6 0 3 p w m 1 o u t m i c r o c o n t r o l l e r w i t h 3 p w m o u t p u t s b h 1 b l 1 i n v _ b l b b m _ e n + v c c b l o c k ( h a l l ) o r s i n e c o m m u t a t e d b l d c m o t o r d i g o u t / h i - z 2 k 2 t m c 6 0 3 m i c r o c o n t r o l l e r w i t h 1 p w m o u t p u t b h 1 b l 1 e n a b l e / e r r _ o u t b b m _ e n + v c c b l o c k ( h a l l f x ) c o m m u t a t e d b l d c m o t o r d i g o u t d i g o u t p w m 1 o u t 2 k 2 d i g i n mosfet gate charge vs . switching event q g C total gate charge ( nc ) v g s C g a t e t o s o u r c e v o l t a g e ( v ) 10 8 6 4 2 0 0 5 10 15 20 25 v d s C d r a i n t o s o u r c e v o l t a g e ( v ) 25 20 15 10 5 0 v m q miller
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 14 copyright ? 2008 trinamic motion control gmbh & co. kg the slope time t slope can be calculated as follows: whereas q miller is the charge the power transistor needs for the switch ing event, and i gate is the driver current setting of the TMC603. taking into account, that a slow switching event means high power dissipation during switching, and, on the other side a fast switching event can cause emv problems, the desired slope will be in some ratio to the switching (chopper) frequency of the system. the chopper frequency is typically slightly outside the audibl e range, i.e. 18 khz to 40 khz. the lower limit for the slope is dictated by the reverse recovery time of the mosfet internal diodes, unless additional schottky diodes are used in parallel to the mosfets source - drain diode . thus, for most applications a switching time between 100ns and 7 5 0ns is chosen. t he required slope control resistor r slp can be calculated as follows: example: a circuit using the transistor from the diagram above shall be designed for a slope time of 2 00ns. the miller charge of the transistor i s about 6nc. the nearest available resistor value i s 330 k . it sets the gate driver current to roughly 30 ma. this is well within the minimum and maximum r slp resistor limits. 5.2.5 reverse capacity (qgd) p rotect ion th e principle of slope control often is realized by gate series resistors with competitors products, but, as latest mosfet gene rations have a fairly high gate - drain charge (q gd ), this approach is critical for safe bridge operation. if the gate is not hel d in the off state with a low resistance, a sudden raise of the voltage at the drain (e.g. when switching on the complementary transistor) could cause the gate to be pulled high via the mosfets gate drain capacitance. this would switch on the transistor an d lead to a bridge short circuit. the TMC603 provides for safe and reliable slope controlled operation by switching on a low resistance gate protection transistor (see figure). figure 9 : qgd protecte d driver stage s d g i o f f i o n v g a t e e x t e r n a l m o s f e t o n o f f s l o p e c o n t r o l l e d f u l l , s a f e o f f q g d q g s t m c 6 0 3 q g d p r o t e c t e d d r i v e r s t a g e
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 15 copyright ? 2008 trinamic motion control gmbh & co. kg 5.2.6 considerations for qgd protection this chapter gives the background understanding to ensure a safe operation for mosfets with a gate - drain (miller) charge q gd substantially larger than the gate - source charge q gs . in order to guarantee a sa fe operation of the q gd protection, it is important to spend a few thoughts on the slope control setting. please check your transistors data sheet for the gate - source charge q gs and the gate - drain charge q gd (miller charge). in order to turn on the mosfet , first the gate - source charge needs to be charged to the transistors gate. now, the transistor conducts and switching starts. during the switching event, the additional q gd needs to be charged to the gate in order to complete the switching event. whereve r q gd is larger than q gs , a switching event of the complementary mosfet may force the gate of the switched off mosfet to a voltage above the gate threshold voltage. for these mosfets the q gd protection ensures a reliable operation, as long as the slopes ar e not set too fast. calculating the maximum slope setting for high q gd mosfets: taking into account effects of the mosfet bulk diode (compare chapter 5.2.7 ), t he maximum slope of a mosfet bridge will be around the double slo pe as calculated from the miller charge and the gate current . based on this, we can estimate the current required to hold the mosfet safely switched off: during the bridge switching period, the driver must be able to discharge the difference of q gd and q g s while maintaining a gate voltage below the threshold voltage. therefore thus the minimum value required for i off qgd can be calculated: wher e i on is the gate current set via r slp , and i off qgd is the q gd protection gate current. the low side driver has a lower q gd protection current capability than the high side driver, thus we need to check the low side. with its r lsoff qgd of roughly 15 ohm, the TMC603 can keep the gate voltage to a level of: now we just need to check u goff against the mosfets output characteristics, to make sure, that no significant amount of drain current can flow. example: a mosfet, where qgd is 3 times larger than qgs is dri ven with 100ma gate current. the TMC603 thus can keep the gate voltage level to a maximum voltage of u goff = 133ma * 15? = 2v this is sufficient to keep the mosfet safely off.
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 16 copyright ? 2008 trinamic motion control gmbh & co. kg note : do not add gate series resistors to your mosfets! this would eliminate the effect of the q gd protection. gate series resistors of a few ohms only may make sense, when paralleling multiple mosfets in order to avoid parasitic oscillations due to interconnection inductivities. 5.2.7 effects of the mosfet bulk diode whenever inductive loads are driven, the inductivity will try to sustain current when current becomes switched off. during bridge switching event s, it is important to ensure brea k - before - make operation, e.g . one mosfet becomes switches off, before the opposite mosfet is switched on. depending on the actual direction of the current, this results in a short moment of a few 100 nanoseconds, where the current flowing through the inductive load forces the bridge output below the lower supply rail or above the upper supply rail. the respective mosfet bulk diode in this case takes over the current. the diode saturates at about - 1.2v. but t he bulk diode is not an optimum device. it typically has reverse recovery time of a few ten to several 100ns and a reverse recovery charge in the range of some 100nc or more. assuming, that the bulk diode of the switching off mosfet takes over the current, the complementary mosfet sees the sum of the coil current and the instantaneo us current needed to recover the bulk diode when trying to switch on. the reverse recovery current may even be higher than the coil current itself! as a result, a number of very quick oscillations on the output appear, whenever the bulk diode leaves the re verse recovery area, because up to the half load current becomes switched off in a short moment. the effect becomes visible as an oscillation due to the parasitic inductivities of the pcb traces and interconnections. while this is normal, it adds high curr ent spikes, some amount of dynamic power dissipation and high frequency electromagnetic emission. due to its high frequency, t he ringing of this current can also be seen on the gate drives and thus can be easily mistaken as a gate driving problem. figure 10 : effect of bulk diode recovery a further conclusion from this discussion: do not set the bridge sl ope time higher than or near to the reverse recovery time of the mosfets, as the parasitic current spikes will multiply the instantaneous current across the bridge. a plausible time is a factor of three or more for the slope time. if this cannot be tolerat ed please see the discussion on adding schottky diodes. 5.2.8 adding schottky diodes across the mosfet bulk diodes in order to avoid effects of bulk diode reverse recovery, choose a fast recovery switching mosfet. the mosfet transistors can also be bridged by a schottky diode, which has a substantially faster reverse recovery time. th is schottky diode needs to be chosen in a way that it can take over the full bridge current for a short moment of time only. during this time, the forward voltage needs to be lower than the mosfets forward voltage. a small 5a diode like the sk56 can take over a current of 20a at a forward voltage of roughly 0.8v. u b m x - 1 . 2 v v v m p h a s e o f s w i t c h i n g e v e n t i h s 0 a n o r m a l s l o p e l s b u l k d i o d e c o n d u c t i n g i o u t o v e r s h o o t + r i n g i n g i l s b u l k 0 v h s c u r r . r i s e u p t o i o u t s w i t c h i n g c o m p l e t e h s s t a r t s c o n d u c t i n g 0 a h s t a k e s o v e r o u t p u t c u r r e n t l s b u l k r e v e r s e r e c o v e r y i o u t - i o u t
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 17 copyright ? 2008 trinamic motion control gmbh & co. kg figure 11 : parallel schottky diode avoids current spikes due to bulk diode recovery 5.2.9 short to gnd detection an overload condition of the high side mosfet (short to gnd) is detected by the TMC603, by monitoring the bm voltage during high side on time. under normal conditions, the high side power mosfet reaches the bridge supply voltage minus a small voltage drop during on time. if the bridge is overloaded, the voltage cannot rise to the detection level within a limited time, de fined by an external resistor. upon detection of an error, the error output is activated. by directly tying it to the enable input, the chip becomes disabled upon detection of a short condition and the error flip flop becomes set. a variation of the short to gnd detection delay allows adaptation to the slope control, and modification of the sensitivity of the detection during power up. figure 12 : timing of the short to gnd detector pin comments rs2g the resistor connected to this pin controls the delay between switching on the high side mosfet and the short to gnd check. a 2 0 a current out of this pi n (resistor value of 22 0 k to gnd) results in a 500 ns delay, a lower current gives a longer delay. disconnecting the pin disables the function. keep interconnection between ic and resistor short, to avoid stray capacitance to adjacent signal traces of mo dulating the set current. resistor range: 47 k to 1 m 5.2.10 error logic the TMC603 has three different sources for signaling an error: undervoltage of the low side supply undervoltage of the charge pump short to gnd detector upon any of these events the erro r output is pulled low. after a short to gnd detector event, t he error output remains active, until it becomes cleared by the clr_err. by tying the error output to the enable input, the TMC603 automatically switches off the bridges upon an error. the enabl e input then should be driven via an open collector input plus pull - up resistor, or via a resistor. h s 1 b m 1 l s 1 2 2 0 r g n d p + v m z 1 2 v m o t o r short detection valid area bmx bhx v vm - v bms 2 g 0 v 0 v v vm t s 2 g bmx voltage monitored short to gnd monitor phase / errout , enable 0 v t s 2 g short detected delay delay inactive inactive short to gnd detected driver off via enable pin
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 18 copyright ? 2008 trinamic motion control gmbh & co. kg figure 13 : error logic pin comments /err_out error output (open drain). signals undervoltage or overcurrent. tie to enable for direct self protection of the driver . the internal error condition generator has a higher priority than the clr_err input, i.e. the erro r condition can not be cleared, as long as it is persistent. clr_err reset of error flip - flop (active high). clears error condition . the error condition should at least be cleared once after ic power on. enable enables the power drivers (low: all mosfets become actively switched off) gnd d / err _ out short to gnd 1 clr _ err d undervoltage vls enable d feedback connection for automatic self - protection 100 k + v cc pull - up resistor can be internal to microcontroller drive with open drain output , if feedback is provided undervoltage vcp short to gnd 2 short to gnd 3 s r q q s : priority tmc 603 error logic
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 19 copyright ? 2008 trinamic motion control gmbh & co. kg 5.3 current measurement amplifiers the TMC603 amplifies the voltage drop in the three lower mosfet transistors in order to allow current measurement without the requirement for c urrent sense (shunt) resistors. this saves c ost and board space, as well as the additional power dissipation in the shunt resistors. however, additional shunt resistors can be added, e.g. source resistors for each lower mosfet or a common shunt resistor in the bridge foot point, in order to increase voltage drop and to have a more exact measurement. the TMC603 curx outputs deliver a signal centered to 1/3 of the 5v vcc supply. this allows measurement of both, negative and positive signals, while staying compatible to a 3.3v microcontroller. the cur rent amplifier is an inverting type. figure 14 : schematic of current measurement amplifier s pin co mments curx output of current measurement amplifier . the output signal is centered to 1/3 vcc. sense_hi switches current amplifiers to high sensitivity (high level). control by processor to get best sensitivity and resolution for measurement. samplex op tional external control for current measurement sample/hold stage. set to positive level, if unused the voltage drop over the mosfet is calculated as follows: whereas x is the adc output value, x 0 is the adc output value at zero current (e.g. 85 for an 8 bit adc with 5v reference voltage), adc max is the range of the adc (e.g. 256 for an 8 bit adc) , v adcref is the reference voltage of the adc and a cur is the currently selected amplification (absolute value) of the TMC603. with this, the motor current can be calculated using the on resistance r dson (at 10v) of the mosfet: 5.3.1 current measurement timing current measurement is self - timed, in order to only provide valid output voltages. sampling is active during the low side on time. the sampling is delayed by an internal time delay, in order t o avoid sampling of instable values during settling time of the bridge current and amplifiers. thus, a minimum on time is required in order to get a current measurement. the output curx reflects the current during the measurement. t he last value is held in a track and hold circuit as soon as the low side transistor switches off. track & hold stage samplex d a automatic sample point delay bmx blx d curx a swc d sense _ hi r r r + v cc 1 / 3 vcc autozero amplify 4 . 5 x or 18 x add 1 / 3 vcc offset
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 20 copyright ? 2008 trinamic motion control gmbh & co. kg figure 15 : timing of the current measurement the samplex pins will normally be unused and can be tied to vcc. for advanced applications, where a precise setting of the current sampling point is desired, e.g. centered to the on - time, samplex pins can be deactivated at the desired p oint of time, enabling the hold stage. 5.3.2 auto zero cycle t he current measurement amplifiers do an automatic zero cycle during the off time of the low side mosfets. the zero offset is stored in internal capacitors. this requires switching off the low side a t least once, before the first measurement is possible, and on a cyclic basis, to avoid drifting away of the zero reference. this normally is satisfied by the chopper cycle. if commutation becomes stopped, e.g. due to motor stand still, the respective phas e current measurement could drift away. after the first switching off and on of the low side, the measurement becomes valid again. 5.3.3 measurement depending on chopper cycle if the low side on - time on one phase t blhicurx is too low, a current measurement is n ot possible. the TMC603 automatically does not sample the current if the minimum low side - on time is not met. this condition can arise in normal operation, e.g. due to the commutation angle defined by a sine commutation chopper scheme. the respective curx output then does not reflect the phase current . thus , the curx output of a phase should be ignored , if the on - time falls below the minimum low side on - time for current measurement (please refer to maximum limit) . the correct current value can easily be cal culated from the difference of the remaining t w o current measurements. 5.3.4 compensating for offset voltages in order to measure low current values precisely, the zero value (x 0 ) of 1/3 vcc can be measured via the adc , rather than being hard coded into the m easurement software. this is possible by doing a first current measurement during motor stand - still, with no current flowing in the motor coils, e.g. during a test phase of the unit. the resulting value can be stored and used as zero reference. however, th e influence of offset voltages can be minimized, by using the high sensitivity setting of the amplifiers for low currents, and switching to low sensitivity for higher currents. 5.3.5 getting a precise current value the on - resistance of a mosfet has a temperatu re co - efficient, which should not be ignored. thus, the temperature of the mosfets must be measure d , e.g. using an ntc resistor, in order to compensate for the variation. also, the initial rdson depends upon fabrication tolerance of the mosfets . if exact m easurement is desired, an adjustment should be done during initial testing of each product. for applications, where an adjustment is not possible, the use of at least one additional shunt resistor in the common ground line of all three half bridges provide s a stable current measurement base, which allows in - system adjustment of the relative mosfet resistances. further , the TMC603 measurement amplification is slightly different for positive and negative currents. sam - plex blx c o n t r o l s i g n a l s bmx curx 0 v 0 v 0 v 0 v internal sample control external control b r i d g e v o l t a g e d r o p v vcc / 3 0 . 25 v - 0 . 25 v c u r r e n t s e n s e o u t v vm t blhicurx t blhicurx curx tracking - bmx hold curx tracking - bmx hold hold ( undef .) phase
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 21 copyright ? 2008 trinamic motion control gmbh & co. kg 5.4 h all fx sensorless commutation h all fx provide s emulated h all sensor signals. the emulated hall sensor signals are available without a phase shift and there is no error - prone pll necessary, like with many other systems, nor is the knowledge of special motor parameters required. since it is based on th e motors back - emf, a minimum motor velocity is required to get a valid signal. therefore, the motor needs to be started without feedback, until the velocity is hig h enough to generate a reliable h all fx signal. figure 16 : h allfx block diagram and timing a switched capacitor filter for each coil supplies the measured effective coil voltages. its fil ter frequency can be adapted to the chopper frequency and the desired maximum motor velocity. an induction pulse suppressor unit gates the commutation spikes which result from the inductive behavior of the motor coils after switching off the current . the g ating time can be adapted by an external capacitor to fit the motor inductivity and its (maximum) velocity. pin comments sp_sup a capacitor attached to this pin sets the spike suppression time. this pin charge s the capacitor via an internal current sourc e. if more exact timing is required, an external 47k pull - up resistor to vcc can be used in parallel to the internal current source . the capacitor becomes discharged upon each valid commutation. the capacitor can optionally be left away, and the suppressio n can be done in software. filtx these pins provide the filtered coil voltages. for most applications this will be of no use, except when an external back - emf commutation is realized, e.g. using a microcontroller with adc inputs. because of the high outpu t resistance and low current capability of these pins, it is advised to add an external capacitor of a few hundred pico farad up to a few n anofarad to gnd, if the signals are to be used. this prevents noise caused by capacitance to adjacent signal traces to disturb the signal. hx emulate d hall sensor output signal of h all fx block. scclk an external clock controls the corner frequency of the switched capacitor filter. a 1.25 mhz clock gives a filter bandwidth of 3khz. a lower clock frequency may be bet ter f or lower motor velocities . 5.4.1 adjusting the h all fx s pike suppression time h all fx needs two minimum motor - and application - specific adjustments : the switched capacitor clock frequency and the spike suppression time should be adapted . both can easily be deduc ted from basic application parameters and are not very critical . the scclk frequency should be matched to the chopper frequency of the system and the maximum motor velocity. the spike suppression time needs to be adapted to the desired maximum motor veloci ty. low pass lpu low pass lpv low pass lpw psg e 1 e 3 e 2 u lp v lp w lp position signal generation ( psg ) induction pulse supressor ( ips ) ips e 1 e 2 e 3 u v w h 1 h 2 h 3 d bm 1 h 1 d h 2 d h 3 bm 2 bm 3 a sp _ sup 30 k a 30 k a 30 k a f i l t 3 f i l t 2 f i l t 1 c sup d s c _ c l k switch cap filters
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 22 copyright ? 2008 trinamic motion control gmbh & co. kg calculating the commutation frequency f c om of the motor: s rpm is the rotation velocity in rpm n pole is the pole count of the actual motor, or the double of the number of pole pairs the spike suppression time can be chosen as high, as the commutation frequency require d for maximum motor velocity allows. as a thumb rule, we take half of this time to have enough spare. example: giv en a 4 pole motor operating at 4000 rpm: c sup = 6.25nf . the nearest value is 6.8nf. 5.4.2 adjusting the h all fx filter frequency the filter block needs to separate the motors back emf from the chopper pulses. thus, the target is, to filter away as much commuta tion noise as possible, while maintaining as much of the back emf signal as possible. therefore, we need to find a cut - off frequency in between the chopper frequency and the electrical frequency of the motor. since we do not want to change the frequency wi thin the application, we use the nominal or maximum motor velocity to calculate its electrical frequency. the chopper frequency is given by the system, typically about 20 khz. the electrical frequency of the motor is: since the filter has a logarithmic behavior, as a thumb rule we can make a logarithmic mean - value as follows: with the cut - off frequency being about 1/390 of the switched capacitor clock frequency f scclk the following results as a thumb rule: the result shall be checked against minimum limit of 250 khz and maximum limit of 4 mhz, however, the actual frequency is q uite uncritical and can be varied in a wide range. example: given a 4 pole motor operating at 4000 rpm with a 20 khz chopper frequency: f el = 133 hz f cutoff = 1.6 khz f scclk = 0.64 mhz the result is well within the limits, however, the frequency in a pr actical application can be chosen between 300 khz and 1.5 mhz. 5.4.3 block commutation chopper scheme for h all fx h all fx works perfectly with nearly every motor. you can use a standard block commutation scheme, but the chopper must fulfill the following: the co ils must be open for some percentage of the chopper period, in order to allow the back - emf of the motor to influence the coil voltages. the motor direction
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 23 copyright ? 2008 trinamic motion control gmbh & co. kg is determined by the start - up scheme, since the h all fx signals depend on the direction. thus, the sa me commutation scheme is used for turn right and turn left! only a single commutation table is required. figure 17 : h allfx based commutation a chopper scheme fulfilling the desired coil open time per chopper period is shown here: both, the high side driver and the low side driver are chopped with the same signal . the coil open time automatically is inve rted to the duty cycle. in a practical application, the motor can run with a duty cycle of 15% to 25% (minimum motor velocity at low load) up to 90% to 95% (maximum motor velocity). the exact values depend on the actual motor. with a lower duty cycle the m otor would not start, or back emf would be too small to yield a valid h all fx signal. with a higher duty cycle, the back emf would not be visible at the coil voltages, because the coils would be connected to gnd or vm nearly the whole time. the minimum resu lting coil open time thus is 5% to 10%. this simple chopper scheme automatically gives a longer measuring time at low velocities, when back emf is lower. the actual borders for the commutation should be checked in the actual application. provide enough hea droom to compensate for variations due to motor load, mechanics and production stray. 5.4.4 start - up sequence for the motor with forced commutation in order t o start the motor running with h all fx , it must reach a minimum velocity. th e microcontroller needs to take care of this by starting the motor in a forward control mode, without feedback C just like a stepper motor. in order to allow a smooth transition to feedback mode, the same chopper scheme should be used as described above. alternatively, the chopper s cheme can be changed a few electrical periods before you switch to h all fx . this allows for example to start - up the motor using a sine commutation, to get a smooth movement also at low motor velocities. in a practical application, only a few percent up to 1 0% of the maximum motor velocity are sufficient for h all fx operation. turn the motor up to a minimum velocity , where you safely get correct h all fx signals. since rotation of the motor can not be measured during this phase, the motor needs to be current co ntrolled, with a current which in every case is high enough to turn the mechanical load. current control can be done by feedback control, or by adapting the duty cycle to the motor characteristics. further, the minimum starting speed and acceleration needs to be set matching the application. for sample code, please see www.trinamic.com . u pon reaching the threshold for h all fx operation, a valid hall signal becomes h 1 h 2 h a l l f x s i g n a l s 0 v 0 v m o t o r t u r n i n g f o r w a r d 3 h a l l v e c t o r h 3 0 v b r i d g e c o n t r o l s i g n a l s ( h i g h a c t i v e ) b h 1 0 v b l 1 0 v b h 2 0 v b l 2 0 v b h 3 0 v b l 3 0 v m o t o r t u r n i n g r e v e r s e 1 5 4 6 2 5 1 3 2 6 4 c h o p p e r o n h i g h s i d e c h o p p e r o n l o w s i d e e x a m p l e : 5 0 % c h o p p e r o n h i g h a n d l o w s i d e s h o w i n g 3 c h o p p e r e v e n t s ( c h o p p e r e v e n t s n o t s h o w n )
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 24 copyright ? 2008 trinamic motion control gmbh & co. kg available and allows checking success of the starting pha se. the turning direction of the start - up sequence automatically determines the dir ection of motor operation with h all fx . you can check velocity and direction, as soon as valid h all fx signals are available. when you experience commutation sequence errors during motor operation, probably motor velocity has dropped below the lower threshold. in this case, the motor could be restarted in forward control mode, or you could switch to forward control mode on the fly.
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 25 copyright ? 2008 trinamic motion control gmbh & co. kg 5.5 power supply the TMC603 integrates a +12v switching regulator for the gate driver supply and a +5v linear regulator for supply of the low voltage circuitry . the switching regulator is designed in a way, that it provides the charge pump voltage by using a vil l ard voltage doubler circuit. it is able to provide enough current to supply a number of digital circuits by adding an additional 3.3v or 5v low voltage linear or switching regulator. if a +5v microcontroller with low current requirement is used, the +5v regulator is sufficient, to also supply t he microcontroller. figure 18 : power supply block pin comments cosc oscillator capacitor for step down re gulator . a 470pf capacity gives 100khz operation. do not leave this pin unconnected. tie to gnd, if oscillator is not used. swout switch regulator transistor output . the output allows driving of a small signal p - channel mosfets as well as pnp small signal transistors 5vout output of internal 5v linear regulator. provided for vcc supply 5.5.1 switching regulator and charge pump the switching regulator has been designed for high stability. it provides an upper duty cycle limit, in order to ensure switching oper ation even at low supply voltage. this allows the combination with a villard voltage doubler. the application schematic shows a number of standard values, however, the coil and oscil lator frequency can be altered: the c hoice of the external switching regu lator transistor depends on the desired load current and the supply voltage. especially for high switching frequencies, a low gate charge mosfet is required. the following table shows an overview of available transistors and indicative operation limits. fo r a higher output current, two transistors can be used in parallel. in this case the switching frequency should be halved, because of the higher gate charge leading to slower switching slopes. + v m s w o u t v l s 5 v l i n e a r r e g u l a t o r 5 v o u t 1 0 0 n f v m v c c g n d 1 0 0 v c p b a v 9 9 ( 7 0 v ) b a s 4 0 - 0 4 w ( 4 0 v ) s s 1 6 t p 0 6 1 0 k o r b s s 8 4 ( o p t . b c 8 5 7 ) l s w v m - 1 2 v / 2 m a d r i v e r c o s c 1 0 0 n ( 2 x ) 2 2 0 n c o s c : 4 7 0 p - > 1 0 0 k h z s m d i n d u c t . 1 h o r 4 r 7 1 0 0 n ( 2 x ) o p t i o n a l s u p p l y f i l t e r c o m p o n e n t s w h e n s u p p l y r i p p l e i s h i g h d u e t o l o w f i l t e r c a p a c i t y f o r t r a n s i s t o r b r i d g e s l s w : 2 2 0 h f o r 1 0 0 k h z v m + 1 0 v c h a r g e p u m p 5 v s u p p l y 1 2 v s u p p l y ( 1 5 0 m a w i t h s e l . t r a n s i s t o r ) 4 7 t a n t a l 2 5 v 1 2 2 0 n 1 6 v t m c 6 0 3 v o l t a g e r e g u l a t o r s t r i a n g l e o s c 1 4 k s r q q r 1 / 5 r r + v c c 4 / 5 r d u t y c y c l e l i m i t r 2 r 1 r + v c c 1 0 r 1 5 0 m v t r i a n g l e 5 / 1 2 v l s s t a r t u p c u r r e n t
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 26 copyright ? 2008 trinamic motion control gmbh & co. kg transistor type manufacturer gate charge (typ.) max. frequency max. voltage max. load current bc857 div. - (bipolar) 100 khz 40v 80 ma bss84 fairchild, nxp 0.9 nc 30 0 khz 50v 120 ma tp0610k vishay 1.3 nc 2 3 0 khz 60v 150 ma nds0605 fairchild 1.8 nc 175 khz 60v 150 ma tp0202k vishay 1 nc 30 0 khz 30v 350 ma for t he catching diode, us e a schottky type with sufficient voltage and current rating. the choice of a high switching frequency allows the use of a smaller and less expensive induc tor as well as a lower capacitance for the villard circuit and the switching re gulator output capacitor. however, the combination of inductor , transistor and switching frequency should be carefully selected and should be adapted to the load current , especially if a high load current is desired . choice of capacitor for the s witching frequency (examples) : c osc frequency f osc inductivity example remark 470 pf 100 khz 220 h 220 pf 175 khz 150 h 100 pf 300 khz 100 h not recommended for v vm < 14v the switcher inductivity shall be chosen in a way, that it can sustain part of the load current between each two switching events. if the inductivity is too low, the current will drop to zero and higher frequency oscillations for the last part of each cycle will result (discontinuous mode) . the required transistor peak current will rise and thus efficiency falls . for a low load current, operation in discontinuous mode is possible. if a high output current is required, a good design value for continuous mode is to target a current ripple in the coil of +/ - 4 0%. to give a coarse hint on the required inductor you can use the following formula for calculating the minimum inductivity required for continuous operation , based on a ripple current which is 100% of the load current : v vm is the supply voltage. for low voltage operation (15v or less), the output voltage sinks from 12v to 0.85*v vm . the formula can be adapted accordingly. i out is the current draw at 12v. for 40% current ripple, you can use roughly the double i nductivity. i f ripple is not critical, you can use a much smaller inductivity, e.g. only 5% to 50 % of the calculated value. but at the same time switching losses will rise and efficiency and current capability sink due to higher losses in the switching tra nsistor . if the TMC603 does not supply additional external circuitry, current draw is very low, about 20ma in normal operation. this would lead to large inductivity values. in this case we recommend going for the values given in the table above in order to optimize coil cost .
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 27 copyright ? 2008 trinamic motion control gmbh & co. kg example: f osc = 175 khz, i out = 0. 2 a , v vm = 48 v : for continuous operation, a 330h or 470h coil would be required. the minimum induct ivity would be around 100h. note : use an inductor, which has a sufficient nominal current rating. keep switching regulator wiring away from sensitive signals. when using an open core inductor, please pay special care to not disturbing sensitive signals. 5.5.2 charge pump the villard voltage doubler circuit relies on the switching regulator generating a square wave at the switching transistor output with a height corresponding to the supply voltage. in order to work properly the load drawn at +12v needs to be higher than the load drawn at the charge pump volt age. this normally is satisfied when the ic is supplied by the step down regulator. for low voltage operation, the charge pump voltage needs to be as high as possible to guarantee a high gate drive voltage , thus, a dual schottky diode shoul d be used for the charge pump in low voltage applications. 5.5.3 supply voltage filtering as with most integrated circuits, ripple on the supply voltage should be minimized in order to guarantee a stable operation and to avoid feedback oscillations via the sup ply voltages. therefore, use a ceramic capacitor of 100nf per supply voltage pin (vm to gnd, vls to gnd and vcc to gnd and vcp to vm). please pay attention to also keep voltage ripple on vcc pin low, especially when the 5v output is used to supply addition al external circuitry. it also is important to make sure, that the resistance of the power supply is low when compared to the load circuit. especially high frequency voltage ripple >1mhz should be suppressed using filter capacitors near the power bridge or near the board power supply. the vm terminal is used, to detect short to gnd situations, thus, it has to correspond to the bridge power supply. in high noise applications, it may make sense to filter vcp supply separately against ripple to gnd. a large lo w esr electrolytic capacitor across the bridge supply (vm to gnd) should also be used, because it effectively suppresses high frequency ripple. this cannot be accomplished with ceramic capacitors. 5.5.4 reverse polarity protection some applications need to be p rotected against a reversed biased power supply, i.e. for automotive applications. a highly efficient reverse polarity protection based on an n channel mosfet can simply be added due to the availability of a charge pump voltage. this type of reverse polari ty protection allows feeding back energy into the supply, and thus is preferable to a pure diode reverse polarity protection. figure 19 : adding a reverse polarity protection 5.5.5 standby with zero power consumption in battery powered application s , a standby function often is desired . it allow s switching the unit on or off without the need for a mecha nical high power switch. in principle , the bridge driver mosfets can switch off the motor completely, but the TMC603 and its switching regulator still need to be switched off in order to reduce current consumption to zero. only a low energy standby power s upply will remain on, in order to wake up the system controller . this standby power supply can be generated by a low current zener diode plus a resistor to the battery voltage, buffered by a capacitor . the example in the schematic uses a p channel mosfet t o switch off power for the TMC603 and any additional + t e r m i n a l v m v c p 1 0 k r e v e r s e p o l a r i t y p o w e r m o s ( i . e . s a m e t y p e a s b r i d g e t r a n s i s t o r s ) b c 8 4 6 1 0 k - t e r m i n a l + v m p r o t e c t e d ( t o b r i d g e )
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 28 copyright ? 2008 trinamic motion control gmbh & co. kg ics which are directly supplied by the battery . in order to safely switch off the motor bridges during standby, the gates of the high side mosfets become actively held in a discharged state using a low c urrent signal mosfet in sot23 package . before entering standby mode, the motor shall be stopped and the TMC603 should be disabled. figure 20 : l ow power standby 5.5.6 low voltage operation down to 9v in low voltage operation, it is important to keep the gate driving voltages as high as possible. the switching regulator for vls thus is not needed and can be left out. tie the pin cosc to gnd. vls becomes directly tied to +vm, which is possible as long as the supply voltage does not exceed 14v (16v peak). however, now a source for the villard voltage double r is missing. a simple solution is to use a cmos 555 timer circuit (e.g. tlc555) oscillating at 250 khz (square wave) to drive the voltage doubler. figure 21 : low voltage operation + v b a t t e r y v m f d c 5 6 1 4 p + v m s w i t c h e d , 3 a m a x . + v m t o b r i d g e , o n l y 2 2 0 n 1 0 0 k 2 7 k e l e c t r o n i c o n s w i t c h e n a b l e t m c 6 0 3 h s x ( o n l y s h o w n f o r o n e h i g h s i d e m o s f e t ) 2 n 7 0 0 2 2 n 7 0 0 2 e n a b l e 1 m s t a n d b y p o w e r s u p p l y 5 - 1 2 v p o w e r s w i t c h s a f e o f f o t h e r b r i d g e s 1 0 n s t d b y s w o u t v l s 5 v o u t 1 0 0 n f v m v c c v c p b a s 4 0 - 0 4 w c o s c 1 0 0 n ( 2 x ) 4 7 0 n s m d i n d u c t . 1 h o r 4 r 7 1 0 0 n ( 2 x ) o p t i o n a l s u p p l y f i l t e r c o m p o n e n t s w h e n s u p p l y r i p p l e i s h i g h d u e t o l o w f i l t e r c a p a c i t y f o r t r a n s i s t o r b r i d g e s v m + 1 0 v c h a r g e p u m p 5 v s u p p l y 1 2 v s u p p l y ( 1 5 0 m a w i t h s e l . t r a n s i s t o r ) 1 1 0 0 n 1 6 v t m c 6 0 3 t l c 5 5 5 v c c r e s e t o u t t r i g g n d c o n t t h r e s d i s c h 2 2 k 1 5 0 p + 9 v . . . 1 4 v
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 29 copyright ? 2008 trinamic motion control gmbh & co. kg 5.6 test output the test output is reserved for manufacturing test. leave open for a normal application . pin comments test output for test voltages. output resistance 25kohm + - 30%. reset: enable(=low); clock: scclk (rising edge). tes t voltage sequence: 0 : 0v 1..3 / 4..6 / 7..9: gate_hs_off, gate_ls_on, ga te_ls_off (driver 1/2/3) 10..14: c urrently unused 1 5: 0v (no further counts: reset for restart)
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 30 copyright ? 2008 trinamic motion control gmbh & co. kg 6 absolute maximum ratings the maximum ratings may not be exceeded under any circums tances. operating the circuit at or near more than one maximum rating at a time for extended periods shall be avoided by application design. parameter symbol min max unit supply voltage v v m - 0.5 50 v supply and bridge voltage max. 20000s 55 v low sid e driver supply voltage v vl s - 0.5 14 v low side driver supply voltage max. 20000s v vl s - 0.5 16 v charge pump voltage (related to gnd) v vc p - 0.5 60 v charge pump voltage max. 20000s 65 v charge pump voltage during power up / down v m - 10 v m +16 v logic supply voltage v vc c - 0.5 6.0 v logic input voltage v i - 0. 5 v cc +0. 5 v analog input voltage v ia - 0. 5 v cc +0. 5 v voltages on driver pins (hsx, lsx, bmx) v drvio - 0.7 0.7 v relative high side driver voltage (v hsx C bmx ) v hsbm - 20 20 v maximum current to / from digital pins and analog low voltage i/os i io +/ - 10 ma 5v regulator continuous output current i 5vout 40 ma 5v regulator short time output current i 5vout 150 ma junction temperature t j - 50 150 c storage temperature t stg - 55 150 c esd - protecti on (human body model, hbm) v esd 1 kv 7 electrical characteristics 7.1 operational range parameter symbol min max unit ambient temperature t a - 40 125 c junction temperature t j - 40 1 4 0 c supply voltage (standard circuit) v v m 10 50 v supply voltage (low vo ltage application : v v ls =v v m ) 9 14 v low side driver supply voltage v vl s 9 1 3 v differential c harge pump voltage measured to vm (v vcp C v vm ) v cp d 8 12 v logic supply voltage v v cc 4.75 5.25 v slope control resistor with v vm <30v r slp 60 500 k slope co ntrol resistor with v vm >30v r slp 10 0 500 k short to gnd control resistor r s2g 47 1000 k output slope t slp 100 1000 ns
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 31 copyright ? 2008 trinamic motion control gmbh & co. kg 7.2 dc characteristics and timing characteristics dc characteristics contain the spread of values guaranteed within the specified supp ly voltage range unless otherwise specified. typical values represent the average value of all parts measured at +25c . temperature variation also causes stray to some values . a device with typical values will not leave min/max range within the full temper ature range. nmos low side driver note 1) dc - characteristics v vc c = 5.0 v, v vl s = 12v parameter symbol conditions min typ max unit gate drive current lsx lo w side switch on i l s on v l s x = 5 v r slp = 6 8 k 125 1 5 0 180 ma gate drive current lsx low side swit ch off i lsoff v lsx = 5v r slp = 68k - 125 - 16 0 - 19 0 ma gate drive current lsx low side switch on i lson v lsx = 5v r slp = 100k 88 105 126 ma gate drive current lsx low side switch off i lsoff v lsx = 5v r slp = 100k - 90 - 1 1 5 - 1 38 ma gate drive current lsx low side switch on i lson v lsx = 5v r slp = 220k 50 ma gate drive current lsx low side switch off i lsoff v lsx = 5v r slp = 220k - 50 ma gate off detector threshold v god v lsx falling 1 v q gd protection resistance after detection of gate off r l soff qgd v lsx = 2 v 1 5 delay ls driver switch on blx to lsx at 50% t lson r slp = 100k c lsx = 100pf 35 70 140 ns delay ls driver switch off blx to lsx at 50% t lso ff r slp = 100k c lsx = 100pf 80 160 320 ns
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 32 copyright ? 2008 trinamic motion control gmbh & co. kg nmos high side driver note 1) dc - characteristics v vc c = 5.0 v, v vl s = 12v, v cpd = 10.5v parameter symbol conditions min typ max unit gate drive current hsx high side switch on i hson v hsx = 5v r slp = 68k 1 12 1 35 180 ma gate drive current hsx high side switch off i hsoff v hsx = v m +5v r slp = 68k - 12 0 - 150 - 180 ma gate drive current hsx high side switch on i hson v hsx = 5v r slp = 100k 75 90 12 0 ma gate drive current hsx high side switch off i hsoff v hsx = v m +5v r slp = 100k - 80 - 10 0 - 12 0 ma gate drive current hsx high side switch on i hson v hsx = 5v r slp = 220k 50 ma gate dr ive current hsx high side switch off i hsoff v hsx = v m +5v r slp = 220k - 50 ma gate off detector threshold high side v hsx - v bm x , bm level high v god v hsx falling v bmx > v gobm 0 v gate off detector threshold high side v bm x , bm level low v gobm v bmx falling 3.5 v q gd protection current after detection of gate off i hsoff qgd v bmx = 24v v hsx = v bmx +2v 30 0 ma delay hs driver switch on bhx to hsx at 50% t hson r slp = 100k v m = 24v c hsx = 100pf 75 150 300 ns delay hs driver switch off bhx to hsx at 50% t h so ff r slp = 100k v m = 24v c hsx = 100pf 60 120 240 ns break - before - make block note 1) timing - characteristics v vm = 48 v, r slp = 100k parameter symbol conditions min typ max unit break - before - make delay lsx off to hsx on t bbmlh measured at 1v gate - source v oltage 160 ns break - before - make delay hsx off to lsx on t bbmhl measured at 1v gate - source voltage 290 ns 1) see timing diagram in figure 6 : bridge driver timing
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 33 copyright ? 2008 trinamic motion control gmbh & co. kg rslp input and rs2g input dc - characteristics v v cc = 5.0 v parameter symbol conditions min typ max unit typical voltage at rslp and rs2g input, depending on the external resistor v rslp v rs 2g r slp = 100 k r s 2g = 100 k 3.8 v r slp = 470 k r s 2g = 470 k 4.0 short to gnd detector dc - char acteristics, timing - characteristics v vm = 24 v parameter symbol conditions min typ max unit short to gnd detection level (v vm C bm ) v bms2g 1 1.5 2.3 v short to gnd detector delay (hs x going active to short detector active / err_out falling) t s2g r s2g = 68k 200 320 450 ns r s2g = 150k 500 750 1000 ns r s2g = 220k 700 1000 1300 ns r s2g = 470k 1400 2000 2600 ns supply current dc - characteristics v vcc = 5.0 v, v vls = 12v, v cpd = 10.5v, r slp = 100k, v vm = 48v parameter symbol conditions min typ m ax unit vm supply current i vm 0.45 0.68 ma vls supply current i vls not including i 5vout 4.6 6.9 ma vcp supply current i vcp 1.6 2.4 ma vcc supply current i vcc 2.9 4.4 ma undervoltage detectors dc - characteristics v vcc = 5.0 v parameter symbol conditions min typ max unit vls undervoltage level v vlsuv 7 7.85 8.5 v vcp undervoltage level (v vcp - v m ) v cpduv v vcp falling 5.8 6.6 v vcp voltage ok level (v vcp - v m ) v vcp rising 7.1 7.8 v vcp undervoltage detector hysteresis 0.5 v
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 34 copyright ? 2008 trinamic motion control gmbh & co. kg switching re gulator / charge pump dc - characteristics v vcc = v 5vout parameter symbol conditions min typ max unit switch output drive current (on) i swout v swout = v vm - 1.5 - 2.2 - 3.0 ma switch output drive current (off) i swout v swout = v vm - 5 v 10 ma switch start - up drive current during vcc undervoltage i swout v swout = v vm v vm = 24v v vls < 2v - 0.4 - 0.8 ma switch output drive voltage (on) v vm - v swout v swout i swout = 0 8 12 15 v switch regulator output voltage v 12vout v vm > 16v 11 12 13.1 v v vlsuv < v vm < 16v 0.85 v vm v oscillator output resistance r cosc t j = 25c 14.1 k lower oscillator threshold voltage v cosc 1/3 v vcc v upper oscillator threshold voltage v cosc 2/3 v vcc v oscillator threshold voltage for maximum duty cycle limit v cosc 6/15 v v cc v maximum duty cycle switch regulator dc swout v vls = 10v f chop = 100khz 63 7 0 77 % chopper frequency nominal f chop c osc = 470pf 70 100 130 khz chopper frequency range (design reference value only) f chop 0 (off) 30 0 khz charge pump voltage (design reference value only) v cpd v vls = 12v i vcp = 1.6ma 10.6 v linear regulator dc - characteristics parameter symbol conditions min typ max unit output voltage v 5vout i 5vout = 10ma t j = 25c 4. 7 5 5.0 5. 2 5 v output resistance r 5vout static load 2 output current capability i 5vout v vls = 12v 100 ma v vls = 8v 60 ma v vls = 6.5v 20 ma
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 35 copyright ? 2008 trinamic motion control gmbh & co. kg digital logic l evel dc - characteristics v vcc = 5.0 v +/ - 10% parameter symbol conditions min typ max unit input voltage low level v inlo - 0.3 0.8 v inp ut voltage high level v inhi 2.0 v vcc +0.3 v output voltage low (h1, h2, h3, err_out) v outlo i outlo = 1ma 0.4 v output voltage high (h1, h2, h3) v outhi i outhi = - 1ma 0.8v vcc v
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 36 copyright ? 2008 trinamic motion control gmbh & co. kg current measurement block dc - characteristics , timing - characteristics v vm = 24 v , v vcc = 5.0 v parameter symbol conditions min typ max unit amplification of voltage v bmx to v curx for p ositive voltage ( v bmx > 0v) a curlo + sense_hi = gnd - 4.3 - 4. 6 - 4. 9 v/v a curhi + sense_hi = vcc - 1 7 - 1 8 . 0 - 19 v/v amplification of v oltag e v bmx to v curx for negative voltage ( v bmx < 0v) a cur lo - sense_hi = gnd - 4. 1 - 4. 4 - 4. 7 v/v a cur hi - sense_hi = vcc - 1 6 - 1 7 . 1 - 1 8 v/v zero current level at curx v 0curx v vcc /3 - 50mv v vcc /3 v vcc /3 +50mv v measurement voltage range at v bmx v bmx sense_hi = gnd - 25 0 300 mv sense_hi = vcc - 8 0 8 0 mv v curx output voltage swing low v curx 0.02 0.1 v v curx output voltage swing high v curx v vcc - 1.2 v vcc - 0.6 v ripple voltage / hold step noise at output note 2 ) v curx v bmx = 0v sense_hi = gnd 17 26 mv v bmx = 0v sense_hi = vcc 50 75 mv minimum low side on time for current measurement (delay from blx going active to curx tracking amplified signal ) t blhi cur x sample x = vcc 3.5 5.3 7.2 s delay from samplex going active to curx tracking amplified signal t smphicurx samplex = vcc t blhicurx / 2 s delay from blx or samplex going inactive to curx hold t blx lo 0 s sample and hold drop during hold period dv curx 0.001 1.6 v/s auto zero drop of current amplifier during sampling period (low side on) dv cur x 0.003 3 v/s minimum initial auto zero period (low side off or samplex low ) after power on t blxlo0 5 s minimum sample period after t blhicurx for a 100% current step t blxhiadd 1 s output current limit at cur x i curx current sourcing 0. 45 ma 2) note on first ics dated before 20/2009: curx outputs are sensible to ripple voltage on vcc pin and frequency below 5mhz. ripple voltage is amplified by 1/3 * set amplification, i.e. factor 1.5 with sense_hi low and factor 6 with sense_hi high. thus, it i s suggested to use 5vout only for vcc supply, if possible, if exact measurements are required.
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 37 copyright ? 2008 trinamic motion control gmbh & co. kg switched capacitor filter 2 nd order dc - characteristics, ac - characteristics v vm = 24 v, v vcc = 5.0 v parameter symbol conditions min typ max unit attenuation of voltage v bmx to v filtx a filtl o v bmx > 0.9v v bmx /15 C filtx 21 30 40 k output current limit at filtx i filtx current sourcing 2 0 a noise voltage on filtx v filtxnoise v bmx = 12v f scclk = 1.25mhz 20 mv 3db bandwidth f cutoff 1/390 f scclk hz f scclk = 1.25mhz 3.0 3.2 3.4 khz f scclk = 2.5mhz 6.4 khz switched ca pacitor filter clock frequency for normal operation f scclk 0.25 4 mhz hall fx unit dc - characteristics, ac - characteristics v vm = 24 v, v vcc = 5.0 v parameter symbol conditions min typ max unit noise voltage of comparators including switched capacitor filter v compnoise v bmx = 12 v 50 150 m v offset voltage of comparators including switched capacitor filter and input attenuation v comp offs v bmx = 12v - 300 0 30 0 mv spike suppression comparator threshold v sp_sup v sp_sup rising 2.0 v vcc /2 2.8 v spike suppr ession capacitor charging current i sp_sup v sp_sup = 1v 15 25 35 a spike suppression capacitor discharging current i sp_sup v sp_sup = 1v - 0.5 - 1 - 1.5 ma dead time for spike suppression t sp_sup c sp_sup = 1nf 60 100 180 s
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 38 copyright ? 2008 trinamic motion control gmbh & co. kg 8 desig ning the application 8.1 choos ing the best fitting power mosfet there is a huge choice of power mosfets available . mosfet technology has been improved dramatically in the last 20 years, and gate drive requirements have shifted from generation to generation . the first generations of mos fets have a comparatively high gate capacity at a moderate rdson. the ir gate - source capacity is two to five times as high as the capacity of the gate - drain junction. these mosfets have a high gate charge and thus require high current gate drive , but they a re easy to use, because internal feedback is low. in the early 2000s new mosfets have emerged, where rdson is much lower, and gate - source capacity has been improved by minimizing structural overlap. thus, the capacitance ratio has shifted, and feedback has become quite high . these mosfets thus are much more critical, and power drives have to actively force the gate off to prevent the bridges from cross - conduction due to feedback from the drain to gate . latest generation mosfets, like the vishay w - fet techno logy, further reduce rdson, while reducing the capacity between the channel and the drain. thus, these mosfets have lowest gate charge, and again, are easier to control than the previous generation of mosfets. when choosing the mosfet, the following point s shall be considered: maximum voltage : choose at least a few volts above your maximum supply voltage, taking into account that the motor can feed back energy when slowing down, and thus the supply voltage can rise. on the other hand, a transistor rated for a higher voltage is more expensi ve and has a higher gate charge (see next chapter). rdson : a low rdson gives low static dissipation, but gate charge and cost increases. take into account, that a good part of the power dissipation results from the swi tching events in a chopped drive system . further, to allow a current measurement, the rdson should be in a range, that the voltage drop can be used for measurement. a voltage drop of 50mv or higher at nominal motor current is a good target. gate charge an d switching speed : the switching speed should be compared to the required chopper frequency. choose the chopper frequency low to reduce dynamic losses. when the application does not require slow, emv optimized switching slopes, choose a low gate charge tr ansistor to reduce dynamic losses. gate threshold voltage: most mosfets have a specified on - resistance at a gate drive voltage of 10v. some mosfets are optimized for direct control from logic ics with 5 or even 3.3v. they provide a low gate threshold vol tage of 1v to 2v. mosfets with higher gate threshold voltage should be preferred, because they are less sensible to effects of the drain gate capacity and cross conduction. packag e, size and cooling requirements cost and availability 8.1.1 calculatin g the mos fet power dissipation the power dissipation in the mosfets has three major components: static losses (p stat ) due to voltage drop, switching losses (p dyn ) due to signal rise and fall times, losses due to diode conduction (p diode ) . the diode power dissipatio n depends on many factors (back emf of the motor, inductivity and motor velocity), and thus is hard to calculate from motor data. normally, it contributes for a few percent to some ten percent of overall power dissipation. other sources for power dissipation are the reverse recovery time of the transistors and the gate drive energy. reverse recovery also cause s current spikes on the bridges. if desired, you can add schottky diodes over the (chopper) transistors to reduce the diode losses and to eliminate current spikes caused by reverse recovery. the following sample calculation assumes a three phase bldc motor operated in block commutation mode with dual sided chopper. at each time, two coils conduct the full motor current (chopped).
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 39 copyright ? 2008 trinamic motion control gmbh & co. kg where i motor is the motor current , e.g. 10a r dson is the on - resistance of the mosfets at a gate voltage of about 10v, e.g. 2 0m ? t duty is the actual duty cycle of the chopper, e.g. 80% = 0.8 v vm is the motor supply voltage, e.g. 24v or 48v f chop is the chopper frequency, e.g. 20khz t slope is the slope (transition) time, e.g. 3 00ns example: with the example data for a 10a motor at 24v , we get the following power dissipation : p stat = 3.2w p dyn24 = 2.88w for comparison: the motor output power is 10a*24v*0.8=192w the dynamic and static dissipation here are in a good ratio, thus the choice of a 20m ? mosfet is good. at 48v, the dynamic power dissipation doubles: p dyn48 =5.76w here, the dynamic losses are higher than the static losses. thus, w e should reduce the slope time. given that the gate capacity would not allow for faster slopes than 300ns, we could go fo r a 30m ? mosfet, which has a lower gate charge and thus allows faster slopes, e.g . 200ns. with these modifications we get a static loss of 4 .8w and a dynamic loss of 3.84w. this in sum is 8.64w, which is slightly less than the 8.96w before. at the same time, syst em cost has decreased due to lower cost mosfets. the loss is still low when compared to a motor power of 384w.
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 40 copyright ? 2008 trinamic motion control gmbh & co. kg 8.2 mosfet examples there is a huge number of mosfets on the market, which can be used in combination with the TMC603. the user choice will depend o n the electrical data (voltage, current, rdson) and on the package and configuration (single / dual). the following table gives a few examples of smd mosfets for different motor currents . the mosfets explicitly are modern types with a low total gate charge . with dual configurations, only three mosfet packages are required to control a bldc motor , but the current which can be reached is significantly lower due to thermal restrictions of the packages . for the actual application, we suggest to calculate stat ic and dynamic power dissipation for a given mosfet, as described in the previous chapter. especially for sine commutation and chopper frequencies above 20khz, transistor s with a gate charge below 100nc should be preferred. t ransistor type manu facturer rdson voltage package & configuration max. motor current (*) total gate charge @10v unit m? v a nc ibp019n06l3 infineon 1.9 60 dpack 30 124 sie812df vishay 2.6 40 polarpa k 30 52 ipp032n06n3 infineon 2.9 60 to220 30 125 irfb3306 international rectifi er 4.2 60 to220 / dpack 30 85 sie816df vishay 7.4 60 polarpak 20 51 sum75n06 - 09l vishay 9.3 60 d2pak (to263) 2 5 47 fdd10an06a0 fairchild 10.5 60 dpak (to252a) 20 28 fdd5353 fairchild 12.3 60 dpak 15 46 si7964dp vishay 23 60 powerpak - so8 (dual) 9.6 43 si4946 vishay 55 60 so8 (dual) 4.5 19 (*) remark: the maximum motor current applicable in a given design depends upon pcb size and layout, since all of these transistors are mainly cooled via the pcb. the data given implies adequate cooling measures tak en by the user, especially for higher current designs. 8.3 driving a dc motor with the TMC603 the TMC603 can also be used for dc motor applications, using a full bridge or a half bridge for motor pwm operation with or without reverse direction operation. for single half bridge applications, all TMC603 gate drivers can be paralleled, taking advantage of the three time increase in gate drive capability. this way a motor current of up to 100a can be driven. the drive system can use the shunt less current sensing capability for best efficiency. a schottky diode across the non - chopped transistor optimizes slopes and electromagnetic emission characteristics (see chapter 5.2.8 ).
TMC603 data sheet (v. 1.0 6 / 26 . mar. 2009) 41 copyright ? 2008 trinamic motion control gmbh & co. kg 9 table of figures figure 1: application block diagram ................................ ................................ ................................ ..... 5 figure 2: pinning / qfn52 package ( top view ) ................................ ................................ ....................... 6 figure 3: application diagram ................................ ................................ ................................ ................ 8 figure 4: three phase bldc driver ................................ ................................ ................................ ..... 10 figu re 5: principle of high - side driver ( pat . fil .) ................................ ................................ ................ 10 figure 6: bridge driver timing ................................ ................................ ................................ ............. 12 figure 7: examples for microco ntroller pwm control ................................ ................................ .. 13 figure 8: mosfet gate charge as available in devi ce data sheet vs . switching event ................. 13 figure 9: qgd protected driver sta ge ................................ ................................ .............................. 14 figure 10: effect of bulk diode recovery ................................ ................................ .......................... 16 figure 11: parallel s cho ttky diode avoids cu rrent spikes due to bulk diode recovery .............. 17 figure 12: timing of the short to gnd detector ................................ ................................ .............. 17 figure 13: error logic ................................ ................................ ................................ ......................... 18 figure 14: schematic of cur rent measurement amp lifiers ................................ ............................... 19 figure 15: timing of the curren t measurement ................................ ................................ ................. 20 figure 16: hall fx block diagram and ti ming ................................ ................................ ....................... 21 figure 17: hall fx based commutation ................................ ................................ ................................ 23 figure 18: power supply block ................................ ................................ ................................ ........... 25 figure 19: adding a reverse pol arity protection ................................ ................................ .............. 27 figure 20: low power standby ................................ ................................ ................................ ............ 28 figure 21: low vol tage operation ................................ ................................ ................................ ...... 28 10 revision history 10.1 documentation revision version author description 0.9 4 bd TMC603 initial release with preliminary electrical data 0.9 6 bd added package dimensions 0.98 bd added microcontroller pwm cont rol examples 0.99 bd added reverse polarity protection and mosfet examples 1.00 bd added low power standby and low voltage operation 1.01 bd removed preliminary indication , modifications in electrical characteristic tables 1.02 bd slightly corrected a few values 1.03 bd added transistor examples and temperature information to tables 1.04 bd slight beautifications / rewording 1.05 bd added mathematical background for qgd protection, discussion on mosfet bulk diode and dc motor application 1.06 bd a dded minimum output voltage swing of current amplifiers table 1 : documentation revisions

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